Controller with constant current limit

ABSTRACT

Methods and apparatuses are disclosed for generating a temperature independent current limit. The value of the temperature independent current limit may be determined based in part on an error signal representative of a difference between an actual output value and a desired output value of a power converter. When the error signal is below a lower threshold voltage, the temperature independent current limit may be set to a first value. When the error signal is above an upper threshold voltage, the temperature independent current limit may be set to a second, higher value. When the error signal is between the lower threshold voltage and the upper threshold voltage, the temperature independent current limit may change linearly with the error signal. The error signal may be adjusted to compensate for changes in the system caused by a change in temperature.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is related to U.S. patent application Ser. No. ______entitle(“Adaptive Biasing” (Attorney Docket No 68660-2000400), filedherewith,

BACKGROUND

1. Field

The present disclosure relates generally to switched mode powerconverters and, more specifically, the present disclosure relates toswitched mode power converters utilizing current mode control.

2. Related Art

Electronic devices use power to operate. Switched mode power convertersare commonly used due to their high efficiency, small size, and lowweight to power many of today's electronics. Conventional wall socketsprovide a high voltage alternating current (ac) power. In a switchedmode power converter, a high voltage ac input is converted to provide aregulated direct current (dc) output through an energy transfer element.In operation, a switch is utilized to provide the desired output byvarying the duty ratio (typically the ratio of the on-time of the switchto the total switching period), varying the switching frequency, orvarying the number of pulses per unit time of the switch in a powerconverter.

The switched mode power converter also includes a controller thattypically provides output regulation by sensing and controlling theoutput in a closed loop. The controller may receive a feedback signalrepresentative of the output and the vary one or more parameters (suchas duty ratio, switching frequency, or the number of pulses per unittime of the switch) in response to the feedback signal to regulate theoutput to a desired quantity. Various modes of control may be utilized.One mode of control is known as pulse width modulation (PWM) peakcurrent mode control. In PWM peak current mode control, the switchremains on until the current in the switch reaches a current limit. Oncethe current limit is reached, the controller turns the switch off forthe remainder of the switching period. In general, a higher currentlimit results in a longer on-time of the switch and a larger duty ratio.

BRIEF DESCRIPTION OF THE DRAWINGS

The above and other aspects, features, and advantages of severalembodiments of the present invention will be more apparent from thefollowing more particular description thereof, presented in conjunctionwith the following drawings.

FIG. 1 shows an example power converter including a controller inaccordance with an embodiment of the present invention.

FIG. 2 is an example of a power converter including a controller thatuses a winding on a coupled inductor to sense the output voltage inaccordance with an embodiment of the present invention.

FIG. 3A is a diagram illustrating an example switching current waveformof a switching power converter operating in continuous conduction mode(CCM) and discontinuous conduction mode (DCM) in accordance with anembodiment of the present invention.

FIG. 3B is a diagram illustrating an example switching current waveformof a switching power converter utilizing pulse width modulation (PWM)current programmed control and operating in DCM in accordance with anembodiment of the present invention.

FIG. 4 is a schematic illustrating an example controller in accordancewith an embodiment of the present invention.

FIG. 5 is a circuit diagram of an example current limit generator inaccordance with an embodiment of the present invention.

FIG. 6 illustrates the relationship between an error voltage and acurrent limit in the example current limit generator of FIG. 5 inaccordance with an embodiment of the present invention.

FIG. 7 illustrates an example process for generating a temperatureindependent current limit in accordance with an embodiment of thepresent invention.

DETAILED DESCRIPTION

In the following description, numerous specific details are set forth inorder to provide a thorough understanding of the present invention. Itwill be apparent, however, to one having ordinary skill in the art thatthe specific detail need not be employed to practice the presentinvention. In other instances, well-known materials or methods have notbeen described in detail in order to avoid obscuring the presentinvention.

Reference throughout this specification to “one embodiment,” “anembodiment,” “one example,” or “an example” means that a particularfeature, structure or characteristic described in connection with theembodiment or example is included in at least one embodiment of thepresent invention. Thus, appearances of the phrases “in one embodiment,”“in an embodiment,” “one example,” or “an example” in various placesthroughout this specification are not necessarily all referring to thesame embodiment or example. Furthermore, the particular features,structures, or characteristics may be combined in any suitablecombinations and/or subcombinations in one or more embodiments orexamples. Particular features, structures, or characteristics may beincluded in an integrated circuit, an electronic circuit, acombinational logic circuit, or other suitable components that providethe described functionality. In addition, it is appreciated that thefigures provided herewith are for explanation purposes to personsordinarily skilled in the art and that the drawings are not necessarilydrawn to scale.

A switched mode power converter includes a controller that typicallyprovides output regulation by sensing and controlling the output in aclosed loop. The controller may receive a feedback signal representativeof the output and the controller may vary one or more parameters (suchas duty ratio, switching frequency, or the number of pulses per unittime of the switch) in response to the feedback signal to regulate theoutput to a desired quantity. Various modes of control may be utilized.One mode of control is known as pulse width modulation (PWM) peakcurrent mode control. In PWM peak current mode control, the switchremains on until the current in the switch reaches a current limit. Oncethe current limit is reached, the controller turns the switch off forthe remainder of the switching period. In general, a higher currentlimit results in a longer on-time of the switch and a larger duty ratio.When the controller utilizes PWM peak current mode control, the accuracyof the current limit is important. However, as will be explained below,the accuracy of the current limit may be susceptible to variations intemperature.

Typically, the received feedback signal is compared to a referencesignal to generate an error signal. In one example, the reference signalmay represent the desired quantity which the output of the powerconverter is regulated to while the error signal is representative ofthe difference between the received feedback signal and the referencesignal. The controller utilizes the error signal to determine the valueof the current limit which will be utilized to control the switch andsubsequently regulate the output. Previous solutions have utilized aresistor to convert the error signal to a current limit As a result, thecurrent limit is also partially determined by the value of the resistor.However, the accuracy of the resistance of the resistor may varydepending on characteristics of the resistor and how the resistor wasmanufactured. For example, a temperature coefficient is associated witha resistor and generally relates the accuracy of the resistance withvariations in temperature. A zero temperature coefficient resistorgenerally refers to a resistor which has a constant resistance withvariations of temperature.

In addition, the controller sets minimum and maximum values for thecurrent limit Previous solutions would set the minimum and maximumvalues by clamping the value of the error signal and then utilizing aresistor to convert the error signal to the minimum and maximum valuesof the current limit. Thus, the accuracy of the current limit ispartially determined by the accuracy of the resistance. For previoussolutions, a zero temperature coefficient resistor was utilized toprovide an accurate current limit (including accurate minimum andmaximum values) over different temperatures. However, zero temperaturecoefficient resistors (such as thin film resistors) are costly. Further,previous solutions would also design the error signal to be temperatureindependent to then provide a temperature independent current limit

In embodiments of the present invention, a control loop is utilized tocompensate for the changes in resistance by subsequently varying thevalue of the error signal. As such, the controller does not need to usea zero temperature coefficient resistor to convert the error signal to acurrent limit However, the control loop compensation breaks down at theminimum and maximum values of the current limit if the error signal isclamped to set the minimum and maximum values. Thus, in embodiments ofthe present invention, a temperature trimmed current reference isutilized to set the minimum and maximum values of the current limitrather than a clamp circuit to clamp the error voltage.

Referring first to FIG. 1, a schematic of an example power converter 100is illustrated including input Y_(IN) 102, an energy transfer element T1104 having a primary winding 106 and a secondary winding 108, a switchS1 110, a clamp circuit 112, a rectifier D1 114, an output capacitor C1116, a load 118, an output quantity U_(O), an output voltage V_(O), anoutput current I_(O), a sense circuit 120, a controller 122, a feedbacksignal U_(FB) 124, a current sense input 126, a drive signal 128, and aswitch current I_(D) 130. Controller 122 further includes a feedbackreference circuit 132, a current limit generator 134, a drive logicblock 136, a feedback reference signal U_(FREF) 138, an error signalU_(ERR) 140, and a current limit I_(LIM) 142. In the illustratedexample, the power converter 100 is shown as a power converter having aflyback topology for explanation purposes. It is appreciated that otherknown topologies and configurations of power converter may also benefitfrom the teachings of the present disclosure.

The power converter 100 provides output power to the load 118 from anunregulated input voltage Y_(IN) 102. In one embodiment, the inputvoltage Y_(IN) 102 is a rectified and filtered ac line voltage. Inanother embodiment, the input voltage Y_(IN) 102 is an unregulated dcinput voltage. The input voltage Y_(IN) 102 is coupled to the energytransfer element T1 104. In some embodiments, the energy transferelement T1 104 may include a coupled inductor. In other embodiments, theenergy transfer element T1 104 may include a transformer. In the exampleof FIG. 1, the energy transfer element T1 104 includes two windings, aprimary winding 106 and secondary winding 108. N_(P) and N_(S) representthe number of turns for the primary winding 106 and secondary winding108, respectively. In the example of FIG. 1, primary winding 106 may beconsidered an input winding and secondary winding 108 may be consideredan output winding. The primary winding 106 is further coupled to theactive switch S1 110, which is then further coupled to the input return111. In addition, the clamp circuit 112 is coupled across the primarywinding 106 of the energy transfer element T1 104.

The secondary winding 108 of the energy transfer element T1 104 iscoupled to the rectifier D1 114. In the example illustrated in FIG. 1,the rectifier D1 114 is exemplified as a diode and the secondary winding108 is coupled to the anode end of the diode. However, in someembodiments, the rectifier D1 114 may be a transistor used as asynchronous rectifier. Both the output capacitor C1 116 and the load 118are coupled to the rectifier D1 114. In the example of FIG. 1, therectifier D1 114 is exemplified as a diode and both the output capacitorC1 116 and the load 118 are coupled to the cathode end of the diode. Anoutput is provided to the load 118 and may be provided as either anoutput voltage V_(O), output current I_(O), or a combination thereof.

In addition, the switched mode power converter 100 further includescircuitry to regulate the output which is exemplified as output quantityU_(O). In general, the output quantity U_(O) is either an output voltageV_(O), output current I_(O), or a combination thereof. A sense circuit120 is coupled to sense the output quantity U_(O). In one embodiment,the sense circuit 120 may sense the output quantity using an additionalwinding on the energy transfer element T1 104. In another embodiment,the sense circuit 120 may sense the output quantity U_(O) from theoutput of the power converter 100.

Controller 122 is further coupled to the sense circuit 120 and includesmultiple terminals. At one terminal, the controller 122 receives afeedback signal U_(FB) 124 from the sense circuit 120. The controller122 further includes terminals for receiving the current sense input 126and outputting the drive signal 128. The current sense input 126provides information associated with the sensed switch current I_(D) 130in switch S1 110. In addition, the controller 122 provides a drivesignal 128 to the switch S1 110 to control various switching parameters.Examples of such parameters may include switching frequency, switchingperiod, duty cycle, or respective on and off times of the switch S1 110.

As illustrated in FIG. 1, the controller 122 includes the feedbackreference circuit 132, current limit generator 134, drive logic block136, feedback reference signal U_(FREF) 138, error signal U_(ERR) 140,and current limit I_(LIM) 142. The feedback reference circuit 132 iscoupled to receive the feedback signal U_(FB) 124 from sense circuit120. In one example, an amplifier may be utilized as the feedbackreference circuit 132, which is sometimes also referred to as an erroramplifier. In one example, feedback reference circuit 132 includes anintegrator that integrates the difference between the reference signalU_(FREF) 138 and the feedback signal U_(FB) 124. In other examples,feedback reference circuit 132 may include a differentiator as well asan integrator to provide desired characteristics for the stability andthe behavior of the power supply in response to changes in input voltageY_(IN) 102 and output current I_(O) as is well known in the art. Thefeedback reference circuit 132 also receives the feedback referencesignal U_(FREF) 138 and outputs the error signal U_(ERR) 140. In thisexample, the error amplifier amplifies the difference between thefeedback reference signal U_(FREF) 138 and the feedback signal U_(FB)124. The output of the feedback reference circuit 132 is provided to thecurrent limit generator 134 as error signal U_(ERR) 140. The output ofthe current limit generator 134 (current limit I_(LIM) 142) is coupledto and received by the drive logic block 136. Drive logic block 136further receives current sense signal 126 and outputs drive signal 128.

In operation, the power converter 100 of FIG. 1 provides output power tothe load 118 from an unregulated input Y_(IN) 102. The power converter100 utilizes the energy transfer element T1 104 to transform the voltagefrom the input Y_(IN) 102 between the primary 106 and secondary 108windings. The clamp circuit 112 is coupled to the primary winding 106 ofthe energy transfer element T1 104 to limit the maximum voltage on theswitch S1 110. Switch S1 110 is opened and closed in response to thedrive signal 128 received from the controller 122. It is generallyunderstood that a switch that is closed may conduct current and isconsidered on, while a switch that is open cannot conduct current and isconsidered off. In the example of FIG. 1, switch S1 110 controls acurrent I_(D) 130 in response to controller 122 to meet a specifiedperformance of the power converter 100. In some embodiments, the switchS1 110 may be a transistor and the controller 122 may include integratedcircuits and/or discrete electrical components. In one embodiment,controller 122 and switch S1 110 are included together in a singleintegrated circuit. In one example, the integrated circuit is amonolithic integrated circuit. In another example, the integratedcircuit is a hybrid integrated circuit.

The operation of switch S1 110 also produces a time varying voltageV_(P) between the ends of primary winding 106. By transformer action, ascaled replica of the voltage V_(P) is produced between the ends ofsecondary winding 108, the scale factor being the ratio that is equal tothe number of turns of secondary winding 108 divided by the number ofturns of primary winding 106. The switching of switch S1 110 alsoproduces a pulsating current at the rectifier D1 114. The current inrectifier D1 114 is filtered by output capacitor C1 116 to produce asubstantially constant output voltage V_(O), output current I_(O), or acombination thereof at the load 118.

The sense circuit 120 senses the output quantity U_(O) to provide thefeedback signal U_(FB) 124 to the controller 122. In the example of FIG.1, the controller 122 also receives the current sense input 126 whichrelays the sensed switch current I_(D) 130 in the switch S1 110. Theswitch current I_(D) 130 may be sensed in a variety of ways, such as,for example, the voltage across a discrete resistor or the voltageacross the transistor when the transistor is conducting.

The controller 122 outputs a drive signal 128 to operate the switch S1110 in response to various system inputs to substantially regulate theoutput quantity U_(O) to the desired value. With the use of the sensecircuit 120 and the controller 122, the output of the switched modepower converter 100 is regulated in a closed loop.

As will be discussed in greater detail below, the current limitgenerator 134 converts the error signal U_(ERR) 140 to a current limitI_(LIM) 142. Along with controller 122, the current limit generator 134provides an accurate current limit I_(LIM) 142 over variations intemperature. While the current limit generator 134 provides an accuratecurrent limit that is temperature independent, some components withinthe current limit generator 134 may be temperature dependent. Thecurrent limit generator 134 also provides an accurate current limitI_(LIM) 142 over temperatures at the minimum and maximum values of thecurrent limit I_(LIM) 142. Current limit I_(LIM) 142 is utilized by thedrive logic 136 to control the operation of switch S1 110. In oneexample, the switch current I_(D) 130 (provided by the current sensesignal 126) is compared to the current limit I_(LIM) 142. Once theswitch current I_(D) 130 reaches the current limit I_(LIM) 142, drivelogic 136 turns switch S1 110 off.

FIG. 2 illustrates an example of a power converter 200 that utilizes awinding on a coupled inductor to sense the output. The power converter200 is illustrated including input V_(IN) 102, an energy transferelement T1 204 having a primary winding 106 and a secondary winding 108,a switch S1 110, a clamp circuit 112, a rectifier D1 114, an outputcapacitor C1 116, a load 118, an output voltage V_(O), an output currentI_(O), a sense circuit 120, a controller 122, a feedback signal U_(FB)124, a current sense input 126, a drive signal 128, and switch currentI_(D) 130. Controller 122 further includes a feedback reference circuit132, a current limit generator 134, a drive logic block 136, a feedbackreference signal U_(FREF) 138, an error signal U_(ERR) 140, and acurrent limit I_(LIM) 142. The sense circuit 120 includes bias winding206. As illustrated in FIG. 2, bias winding 206 is also a part of energytransfer element T1 204.

Power converter 200 is similar to the power converter 100, except thatthe sense circuit 120 utilizes a bias winding 206 on the energy transferelement T1 204 to sense the output voltage. Energy transfer element T1204 may include a coupled inductor or a transformer. FIG. 2 includes theenergy transfer element T1 204 which has a primary winding 106, asecondary winding 108, and a bias winding 206. The sense circuit 120includes the bias winding 206. The sense circuit 120 may further includecircuitry to provide the feedback signal U_(FB) 124 (shown also asfeedback voltage V_(FB)).

In operation, the bias winding 206 produces a voltage V_(B) that isrepresentative of the output voltage V_(O) when rectifier D1 114 onsecondary winding 108 conducts. During the on-time of the switch S1 110,the bias winding 206 produces a voltage V_(B) that is representative ofthe input voltage V_(IN) 102. In another example, bias winding 206 mayalso provide a source of power to the circuits within controller 122.

It is appreciated that many variations are possible in the use of a biaswinding to sense an output voltage and for providing sensing while alsoproviding power to a controller with galvanic isolation. For example, abias winding may apply a rectifier and a capacitor similar to rectifierD1 114 and capacitor C1 116, respectively, to produce a dc bias voltagewhile providing an ac feedback signal from the anode of the rectifier.As such, additional passive components, such as resistors, may be usedon the bias winding to scale the voltage from the bias winding to avalue that is more suitable to be received by controller 122.

In the example of FIG. 2, input voltage V_(IN) 102 is positive withrespect to an input return 111, and output voltage V_(O) 120 is positivewith respect to an output return 117. The example of FIG. 2 showsgalvanic isolation between the input return 111 and the output return117 because input return 111 and output return 117 are designated bydifferent symbols. In other words, a dc voltage applied between inputreturn 111 and output return 117 will produce substantially zerocurrent. Therefore, circuits electrically coupled to the primary winding106 are galvanically isolated from circuits electrically coupled to thesecondary winding 108.

Use of bias winding 206 to sense output voltage V_(O) has the advantagesof providing galvanic isolation between the output voltage V_(O) and thecontroller 122 without the expense of an optocoupler. However, whenusing a winding on coupled inductor 204 to sense output voltage V_(O),the voltage V_(B) at bias winding 206 is representative of outputvoltage V_(O) only when output rectifier D1 114 is conducting. In otherwords, the sense circuit 120 may only sense the output voltage V_(O)during the off-time of the switch S1 110.

The switching current of various modes of operation is illustrated inFIG. 3A. A diagram of an example switching current waveform of theswitched mode power converter of FIG. 1 and FIG. 2 is illustratedincluding switching period T_(S) 304, a switch on-time t_(ON) 306, aswitch off-time t_(OFF) 308, trapezoidal shape 310, and triangular shape312. FIG. 3A illustrates the general waveforms of the switch currentI_(D) 302 with respect to time when the controller is operating in bothcontinuous conduction mode (CCM) and discontinuous conduction mode(DCM). In one example, the general waveforms of switch current I_(D) 302represent the switch current I_(D) 130 illustrated in FIGS. 1 and 2.

During any switching period T_(S) 304, switch S1 110 may conduct inresponse to the drive signal 128 from the controller 122 to regulate theoutput quantity U_(O). The switching period T_(S) 304 may be separatedinto two sections of time: switch on-time t_(ON) 306 and switch off-timet_(OFF) 308. Switch on-time t_(ON) 306 denotes the portion of theswitching period T_(S) 304 where the switch S1 110 may conduct. Switchoff-time t_(OFF) 308 denotes the remaining portion of the switchingperiod T_(S) 304 when the switch S1 110 cannot conduct. The currentwaveform of FIG. 3A illustrates two fundamental modes of operation. Thetrapezoidal shape 310 is characteristic of continuous conduction mode(CCM) whereas the triangular shape 312 is characteristic ofdiscontinuous conduction mode (DCM). During CCM, the switch currentI_(D) 302 is substantially non-zero immediately after the start of theswitch on-time t_(ON) 306. At DCM, the switch current I_(D) 302 issubstantially zero immediately after the beginning of the switch on-timet_(ON) 306. During the switch off-time t_(OFF) 308, the switch currentI_(D) 302 is substantially zero for both CCM and DCM.

Referring next to FIG. 3B, another diagram of an example of a switchingcurrent waveform is illustrated including switch current I_(D) 314,switching period T_(S) 302, switch on-time t_(ON) 306, switch off-timet_(OFF) 308, and a current limit I_(LIM) 316. In one example, FIG. 3Bdemonstrates the general switch current I_(D) 130 of FIGS. 1 and 2 withrespect to time when PWM mode control of the switch S1 110 is utilizedto regulate the output quantity U_(O). In particular, FIG. 3Bdemonstrates the general switch current I_(D) 130 with respect to timewhen PWM peak current control is utilized.

The switch S1 110 conducts at the beginning of each switching periodT_(S) 304. Switch S1 110 conducts until the switch current I_(D) 314reaches the current limit I_(LIM) 316. In one example, control of thecurrent limit I_(LIM) 316 maintains the peak of the switch current I_(D)314 at a value required to regulate the output quantity U_(O). In someembodiments, the length of switching period T_(S) 302 remains constant.In other embodiments, the length of switching period T_(S) 302 may vary.In general, a higher maximum current limit I_(LIM) 316 results in alonger switch on-time t_(ON) 306. The regulation is accomplished by afixed frequency PWM technique known as fixed frequency PWM current modecontrol, fixed frequency PWM current programmed control, and/or PWM peakcurrent mode control.

The term “fixed frequency control” does not necessarily entail that theswitching frequency f_(S) of the switch S1 110 remains unchanged. Theuse of the term “fixed frequency” control indicates that the switchingfrequency f_(S) of the switch is not used as a control variable toregulate the output quantity U_(O). For the example of fixed frequencycurrent mode control, the value of the current limit I_(LIM) 316 isutilized as the control variable to regulate the output quantity U_(O).

Referring next to FIG. 4, a block diagram of an example of controller400 is illustrated including feedback reference circuit 132, currentlimit generator 134, a PWM block 402, and an oscillator 404. The PWMblock 402 includes a comparator 406 and a latch 408. Further illustratedin FIG. 4 are feedback signal U_(FB) 124, current sense signal 126,drive signal 128, feedback reference signal U_(FREF) 138, error signalU_(ERR) 140, current limit I_(LIM) 142, and a clock signal 410.Controller 400 is one example of controller 122 illustrated in FIGS. 1and 2.

The controller 400 further illustrates a controller utilizing PWM peakcurrent mode control. As illustrated, the PWM block 402 is coupled toreceive the current sense signal 126 and output the drive signal 128 toswitch S1 110. As mentioned above, the current sense signal 126 isrepresentative of switch current I_(D) 130. The PWM block 402 is furthercoupled to the current limit generator 134 and receives the currentlimit ILIM 142. Further, the PWM block 402 is coupled to oscillator 404and receives the clock signal 410.

The PWM block 402 further includes comparator 406 and latch 408. In theexample shown, latch 408 is an S-R latch. Comparator 406 receives thecurrent sense signal 126 (and hence the switch current I_(D) 130) at thenon-inverting input and the current limit I_(LIM) 142 at the invertinginput. The output of comparator 406 is then coupled to the reset-inputof latch 408. At the set-input of latch 408, the latch 408 receives theclock signal 410 from oscillator 404. Utilizing the signals received atthe reset-input and set-input, the latch 408 outputs drive signal 128.

In operation, when the clock signal 410 pulses to a logic high valuesignaling the beginning of a switching period T_(S), the output of thelatch 408 transitions from a logic low value to a logic high value andthe drive signal 128 turns the switch S1 110 on. Clock signal 410quickly falls to a logic low value and the output of the latch 408remains at the logic high value. The drive signal 128 (in other words,the output of the latch 408) may transition to a logic low value whenthe value of the current sense signal 126 (i.e. the switch current I_(D)130) reaches the value of the current limit provided by the currentlimit signal I_(LIM) 142. When the switch current I_(D) 130 reaches thecurrent limit I_(LIM) 142, the output of comparator 406 transitions to alogic high value. When the latch 408 receives a logic high value at thereset-input, the output of the latch 408 transitions to a logic lowvalue and the switch S1 110 is turned off. Since the current limitI_(LIM) 142 is partially utilized to control switch S1 110, it isimportant to provide an accurate current limit to regulate output of thepower converter.

FIG. 5 illustrates a circuit diagram of an example current limitgenerator 500. Current limit generator 500 is configured to generate anadjustable temperature independent current limit having a value thatchanges based on the voltage of an error signal that is representativeof a difference between an actual output and a desired output of a powerconverter. Current limit generator 500 may be included within acontroller similar or identical to controller 122 and/or controller 400.

Current limit generator 500 includes supply voltage V_(SUPPLY) 504,temperature independent current source 506, operational amplifiers 516and 534, transistors 508, 510, 518, 520, 522, 530, and 532, and resistor528. Operational amplifiers 516 and 534 with respective transistors 518and 530 are configured as voltage followers, such that the voltage atthe inverting input of each amplifier is substantially equal to thevoltage at the non-inverting input. In the illustrated example, currentlimit generator 500 includes a reference current circuit includingtemperature independent current source 506 and transistors 508 and 510for generating a constant reference current 514. In some examples,temperature independent current source 506 generates a constanttemperature independent current 512 having a constant reference currentI_(REF) that may be conducted through diode connected transistor 508.The gate of transistor 510 is coupled to the gate of transistor 508,thereby coupling transistor 510 to transistor 508 as a current mirror.In the example shown, the ratio of channel width to channel length oftransistor 510 may be the same, or at least substantially the same, asthe ratio of channel width to channel length of transistor 508. However,the ratios may be any value. Thus, transistor 510 may conduct a constantreference current 514 equal, or at least substantially equal, to theconstant current 512 generated by current source 506.

Current limit generator 500 further includes a comparison circuitincluding transistors 518, 530, and 532, operational amplifiers 516 and534, and resistor 528. In some examples, the comparison circuitgenerates an adjustable reference current 540 based on a comparisonbetween the error voltage V_(ERROR) 140 (one example of error signalU_(ERR) 140) and the reference signal V_(LREF) 536. In particular, thecomparison circuit generates an adjustable reference current 540 basedon the difference between the error voltage V_(ERROR) 140 and thereference signal V_(LREF) 536. In the illustrated example, the gate oftransistor 532 is coupled to the gate of transistor 508, couplingtransistor 532 and transistor 508 as a current mirror. However, unliketransistor 510, the ratio of channel width to channel length oftransistor 532 may be “N” times that of transistor 508. Thus, transistor532 may conduct a current 538 equal to (N×I_(REF)).

Transistor 532 is also coupled to resistor 528, transistor 530, andtransistor 518 such that a current in transistor 532 is the same as thecurrent in transistor 530, resistor 528, and the transistor 518. Thegates of transistors 530 and 518 are coupled to the outputs ofoperational amplifiers 534 and 516, respectively. Operational amplifier516 receives error voltage V_(ERROR) 140, which may be a voltagerepresentation of error signal U_(ERROR) 140, at its non-inverting inputterminal and is coupled to a first end of resistor 528 at its invertinginput terminal. Operational amplifier 534 receives reference voltageV_(LREF) 536, which represents a lower threshold for error voltageV_(ERROR) 140, at its inverting input terminal and is coupled to asecond end of resistor 528 at its non-inverting input terminal. Whencoupled in this way as voltage followers, operational amplifiers 516 and534 force a current 540 having a value I_(ADJ) equal to(V_(ERROR)−V_(LREF))/(R) in transistor 532 when V_(ERROR) 140 is greaterthan V_(LREF) 536 and less than V_(LREF)+(N×I_(REF)×R). However, sincethe ratio of channel width to channel length of transistor 532 is “N”times that of transistor 508, the maximum value I_(ADJ) of current 540is equal to (N×I_(REF)). Therefore, when V_(ERROR) 140 is greater thanV_(LREF) +(N×I_(REF)×R), operational amplifier 534 and transistor 530are unable to hold the voltage at the non-inverting input of operationalamplifier 534 at the value V_(LREF) 536. As a result, the currentI_(ADJ) 540 in resistor 528 remains constant for values of V_(ERROR) 140greater than V_(LREF)+(N×I_(REF)×R). Additionally, when V_(ERROR) 140 isless than reference voltage V_(LREF) 536, operational amplifier 516 andtransistor 518 are unable to conduct current into resistor 528, sotransistor 532 conducts substantially no current.

In some examples, N may be equal to 3 or more. However, it should beappreciated that other values may be used depending on the referencecurrent I_(REF) and the desired maximum value of temperature independentcurrent limit 142. Additionally, the resistance R of resistor 528 may beselected based on the available voltage (e.g., V_(SUPPLY) 504) and theamount of temperature independent current limit 142 that is desired. Insome examples, resistor 528 can include any type of resistor that has amonotonic variation with current and temperature. In other words, thevalue of the resistor always changes in the same direction withincreasing current and increasing temperature. In other examples,resistor 528 may include a temperature independent resistor. In furtherexamples, resistor 528 need not be monotonic.

Current limit generator 500 further includes an output circuit includingtransistors 520 and 522. In some examples, the output circuit may outputtemperature independent current limit 142 based at least in part on theconstant reference current 514 generated by the reference currentcircuit and the adjustable reference current 540 generated by thecomparison circuit. In the illustrated example, diode connectedtransistor 520 is coupled to supply voltage V_(SUPPLY) 504, transistor510, transistor 518, and transistor 522. In this way, the current intransistor 520 is equal to the current in transistor 510 (current 514)plus the current in transistor 518, transistor 530, resistor 528, andtransistor 532 (current 540). As mentioned above, the current 514 intransistor 510 has a value equal to the constant reference currentI_(REF), while the current 540 in transistor 518, transistor 530,resistor 528, and transistor 532 may have an adjustable value I_(ADJ)between zero and (N×I_(REF)). Thus, the current in transistor 520 has arange between I_(REF) and (I_(REF)+(N×I_(REF))) amperes (A).

In the output circuit, the gate of transistor 522 is coupled to the gateof transistor 520, thereby coupling transistor 522 to transistor 520 asa current mirror. The channel dimensions of transistor 522 may be thesame, or at least substantially the same, as the channel dimensions oftransistor 520. Thus, transistor 522 may output a temperatureindependent current limit 142 having a value I_(LIM) that is equal, orat least substantially equal (e.g., within 5% or less), to the currentin transistor 520.

In this configuration, current limit generator 500 may generate acurrent limit 142 that is independent of temperature. A control loopformed by controller 122, the output of the power converter, and sensecircuit 120 may compensate for changes in the resistance of resistor 528due to changes in temperature. Specifically, referring simultaneously toFIGS. 1, 2, 4, and 5, if the resistance R of resistor 528 increases(e.g., due to a change in temperature), then the current limit 142 willmomentarily decrease (due to the adjustable current 540((V_(ERROR)−V_(LREF))/(R)) decreasing), thereby reducing the on-time ofswitch 110 and providing less power to the output. This may result in alower output voltage V_(O), thereby increasing the difference betweenthe output voltage (represented by U_(FB) 124 shown in FIGS. 1, 2, and4) and the feedback reference voltage V_(FREF) 138. As a result, theerror voltage 140 increases. Since the current limit is proportional toV_(ERROR)/R, the error voltage V_(ERROR) 140 increases and compensatesfor the increase in the resistance R to return the current limit 142 toits value before the change in temperature caused the change in theresistance R.

Conversely, if the resistance R of resistor 528 decreases (e.g., due toa change in temperature), then the current limit 142 will momentarilyincrease (due to the adjustable current 240 ((V_(ERROR)−V_(LREF))/(R))increasing), thereby increasing the on-time of switch 110 and providingmore power to the output. This may result in a higher output voltageV_(O), thereby decreasing the difference between the output voltage(represented by U_(FB) 124 shown in FIGS. 1, 2, and 4) and the feedbackreference voltage V_(FREF) 138. As a result, the error voltage 140decreases. Since the current limit is proportional to V_(ERROR)/R, theerror voltage V_(ERROR) 140 decreases and compensates for the decreasein the resistance R to return the current limit 142 to its value beforethe change in temperature caused the change in the resistance R.

While the feedback loop may adjust the error voltage VERROR 140 tocompensate for changes in resistance R, the change may not adverselyaffect the controller partially because the gain of the error amplifier132 is high (e.g., 3000 or more). In some embodiments, the change in thefeedback voltage VFB with respect to the feedback reference signal UFREF138 may be small. Specifically, a small change in the feedback voltageVFB with respect to the feedback reference signal UFREF 138 may resultin a relatively large change in the error voltage VERROR 140 due in partto the gain of the error amplifier 132.

By generating a current limit 142 in this way, resistor 528 need not bea temperature independent resistor (otherwise known as a zero TCresistor). As such, components within the current limit generator 500are temperature dependent. Additionally, as will be discussed in greaterdetail below with respect to FIG. 6, the minimum (I_(REF)) and maximum(I_(REF)+(N×I_(REF))) values of current limit 142 is determined by thevalue of temperature independent current source 506 and the ratio ofchannel width to channel length of transistors 508 and 538. Thisobviates the need to use a voltage clamp on the error voltage V_(ERROR)140 to clamp the current limit 142. As a result, the control loop isallowed to change the voltage of error signal U_(ERROR) 140 tocompensate for changes in R for the entire range of values for thecurrent limit 142. If the error voltage V_(ERROR) 140 were clamped, thecontrol loop would not be able to compensate for the change inresistance because the control loop relies on the ability to change thevoltage of error signal U_(ERROR) 140 to compensate for the change inresistance R. Moreover, the voltage to which the error voltage V_(ERROR)140 would be clamped would also have to change with temperature in thesame way as the resistance R.

FIG. 6 illustrates an example graph illustrating the relationshipbetween an error voltage V_(ERROR) 140 and a temperature independentcurrent limit I_(LIM) 142 over varying temperatures. As shown in FIG. 6,the temperature independent current limit I_(LIM) 142 is equal tominimum current limit I_(MIN) 640 when the error voltage V_(ERROR) 140is equal to or less than V_(LREF) 536. Referring back to FIG. 5, whenthe error voltage V_(ERROR) 140 is equal to or less than referencevoltage V_(LREF) 536, transistors 518 and 530 are off and the valueI_(ADJ) of current 540 is equal, or at least substantially equal, tozero. As a result, the current in transistor 520, and thus, thetemperature independent current limit 142 conducted by transistor 522,is equal to I_(REF) (the constant reference current 514 conducted bytransistor 510) when the error voltage V_(ERROR) 140 is equal to or lessthan reference voltage V_(LREF) 536.

When the error voltage V_(ERROR) 140 is between reference voltageV_(LREF) 536 and upper threshold voltage V_(H) 660, 683, and 685, thetemperature independent current limit I_(LIM) 142 changes proportionally(with a slope 680, 682, and 684 of 1/R) with error voltage V_(ERROR)140. FIG. 6 illustrates the relationship of the error voltage V_(ERROR)140 and a current limit I_(LIM) 142 over varying temperatures. As thetemperature increases, the value of R increases and the slope of thecurrent limit 142 decreases. To illustrate, slope 682 is greater thanslope 680, which in turn is greater than slope 684. Slopes 682, 680, and684 correspond to increasing temperatures and a resultant increase inthe value of the resistance R. In addition, the corresponding upperthreshold voltage V_(H) (683, 660, and 685 respectively) also increaseswith temperature.

Referring back to FIG. 5, as the error voltage V_(ERROR) 140 increasesabove reference voltage V_(LREF) 536, transistor 518, transistor 530,resistor 528, and transistor 532 conduct the current 540 having valueI_(ADJ). The current 540 having value I_(ADJ) linearly increases sinceresistor 528 is a constant value for a given temperature. As a result,the current in transistor 520, and thus, the temperature independentcurrent limit I_(LIM) 142 in transistor 522, is equal to I_(REF) (theconstant reference current 514 in transistor 510) plus the linearlyincreasing current 540 having value I_(ADJ) when the error voltageV_(ERROR) 140 is between reference voltage V_(LREF) 536 and upperthreshold voltage V_(H).

As shown in FIG. 6, the temperature independent current limit I_(LIM)142 is equal to the maximum current limit I_(MAX) 620 ((1+N)×I_(REF))when the error voltage V_(ERROR) 140 is equal to or greater than upperthreshold V_(H). Upper threshold V_(H) corresponds to the value of theerror voltage V_(ERROR) 140 that causes the current 540 having valueI_(ADJ) to be clamped to N times I_(REF). Referring back to FIG. 5 andas discussed above, the temperature independent current limit 142linearly increases as the error voltage V_(ERROR) 140 increases abovereference voltage V_(LREF) 536. However, as the error voltage V_(ERROR)140 increases to upper threshold voltage V_(H), the current 540 reachesits maximum value of (N×I_(REF)) and remains constant, or at leastsubstantially constant, for values of error voltage V_(ERROR) 140greater than V_(H). For values of error voltage V_(ERROR) 140 greaterthan V_(H), operational amplifier 534 and transistor 530 cannot controlthe voltage at the non-inverting input of operational amplifier 534, andthe current in resistor 528 is determined only by the mirrored referencecurrent in transistor 532. As a result, the current in transistor 520,and thus, the temperature independent current limit 142 in transistor522, is equal to reference current I_(REF) (the constant referencecurrent 514 in transistor 510) plus the maximum value (N×I_(REF)) ofcurrent 540 when the error voltage V_(ERROR) 140 is greater than upperthreshold voltage V_(H).

It should be appreciated that the value of upper threshold voltage V_(H)may change with the resistance R and as such, may also change withtemperature. In one example, the upper threshold voltage V_(H) issubstantially equal to the reference voltage V_(LREF) 536 added to theproduct of N, current reference I_(REF), and resistance R; ormathematically: V_(H)=V_(LREF)+(NI_(REF)R). For example, as the value ofR increases (with increasing temperature), the error voltage V_(ERROR)140 required to produce the same current also increases. Conversely, asthe value of R decreases (with decreasing temperature), the errorvoltage V_(ERROR) 140 required to produce the same current alsodecreases. However, the value of the minimum and maximum current limitsI_(MIN) and I_(MAX) are independent of the value of resistance R.

Referring now to FIG. 7, an example process 700 for generating atemperature independent current limit is shown. Process 700, begins atblock 705 and proceeds to block 710, where an output of a powerconverter is sensed. For example, a signal representative of the outputvoltage of a power converter may be sensed using a bias winding similaror identical to bias winding 206. In other examples, other techniquesfor sensing an output may be used.

At block 715, the sensed output may be compared to a reference. Forexample, an amplifier similar or identical to amplifier 132 may be usedto compare the sensed output (e.g., feedback signal U_(FB) 124), whichis representative of the output of the power converter, to a reference(e.g., feedback reference signal U_(FREF) 138), which is representativeof a desired output of the power converter.

At block 720, an error signal may be generated. For example, anamplifier similar or identical to amplifier 132 may be used to generatean error signal (e.g., error signal U_(ERR) 140) representative of adifference between a desired output value (e.g., reference signalU_(FREF) 138) and an actual output value (e.g., feedback signal U_(FB)124).

At block 725, it may be determined whether or not the error signal isless than a lower threshold value. For example, a circuit similar oridentical to current limit generator 500 may be used to compare theerror signal (e.g., error signal U_(ERR) 140) to a lower thresholdvalue, such as V_(LREF). If the error signal is less than the lowerthreshold, the process may proceed to block 730 where the temperatureindependent current limit may be set to a minimum value, after which,the process may return to block 710. For example, the temperatureindependent current limit I_(LIM) may be set to I_(REF) if the errorsignal is less than the lower threshold. However, if the error signal isnot less than the lower threshold, then the process may proceed to block735

At block 735, it may be determined whether or not the error signal isgreater than an upper threshold value. For example, a circuit similar oridentical to current limit generator 500 may be used to determine if theerror signal (e.g., error signal U_(ERR) 140) is greater than an upperthreshold value, such as V_(H) 660, 683, and 685. If the error signal isgreater than the upper threshold, the process may proceed to block 740where the temperature independent current limit I_(LIM) may be set to amaximum value, after which, the process may return to block 710. Forexample, the temperature independent current limit I_(LIM) may be set toI_(REF)+(N×I_(REF)) if the error signal is greater than the upperthreshold V_(H). However, if the error signal is not greater than theupper threshold, then the process may proceed to block 745.

At block 745, the temperature independent current limit may be set to avalue according to the error signal. For example, using a circuitsimilar or identical to current limit generator 500, the temperatureindependent current limit may be set to change proportionally with theerror signal (e.g., error signal 140) as shown in FIG. 6. The processmay then return to block 710.

While the blocks of process 700 have been presented in a particularsequence, it should be appreciated that they may be performed in anyorder and that one or more blocks may be performed at the same time.

The above description of illustrated examples of the present invention,including what is described in the Abstract, are not intended to beexhaustive or to be limitation to the precise forms disclosed. Whilespecific embodiments of, and examples for, the invention are describedherein for illustrative purposes, various equivalent modifications arepossible without departing from the broader spirit and scope of thepresent invention. Indeed, it is appreciated that the specific examplevoltages, currents, frequencies, power range values, times, etc., areprovided for explanation purposes and that other values may also beemployed in other embodiments and examples in accordance with theteachings of the present invention.

These modifications can be made to examples of the invention in light ofthe above detailed description. The terms used in the following claimsshould not be construed to limit the invention to the specificembodiments disclosed in the specification and the claims. Rather, thescope is to be determined entirely by the following claims, which are tobe construed in accordance with established doctrines of claiminterpretation. The present specification and figures are accordingly tobe regarded as illustrative rather than restrictive.

What is claimed is:
 1. A circuit for generating a temperatureindependent current limit to regulate an output of a power converter,the circuit comprising: a reference current circuit operable to generatea constant reference current; a comparison circuit operable to: receivean error signal representative of a difference between an actual outputvalue and a desired output value; receive a reference signalrepresentative of a value of the error signal corresponding to a minimumcurrent limit; and generate an adjustable reference current, wherein avalue of the adjustable reference current is variable based at least inpart on a difference between the error signal and the reference signal;and an output circuit coupled to the reference current circuit and thecomparison circuit, the output circuit operable to generate thetemperature independent current limit based at least in part on theconstant reference current and the adjustable reference current.
 2. Thecircuit of claim 1, wherein a minimum value of the temperatureindependent current limit is set based at least in part on a value ofthe constant reference current, and wherein a maximum value of thetemperature independent current limit is set based at least in part onthe value of the constant reference current and a maximum value of theadjustable reference current.
 3. The circuit of claim 2, wherein themaximum value of the temperature independent current limit is equal toat least four times the value of the minimum value of the temperatureindependent current limit.
 4. The circuit of claim 2, wherein theminimum value of the temperature independent current limit and themaximum value of the temperature independent current limit aretemperature independent values.
 5. The circuit of claim 1, wherein thereference current circuit comprises: a temperature trimmed currentsource operable to generate a constant current; a first transistorcoupled to the temperature trimmed current source, wherein the firsttransistor is coupled to receive the constant current from thetemperature trimmed current source; and a second transistor coupled tothe first transistor as a current mirror, wherein the second transistoris coupled to receive the constant reference current.
 6. The circuit ofclaim 5, wherein a value of the constant reference current issubstantially equal to a value of the constant current.
 7. The circuitof claim 5, wherein the comparison circuit comprises: a third transistorcoupled to the first transistor as a current mirror; a fourth transistorcoupled to the third transistor; a fifth transistor coupled to the thirdtransistor and the fourth transistor; a resistor coupled to the thirdtransistor, the fourth transistor, and the fifth transistor; a firstoperational amplifier operable to receive the reference signal, whereinthe first operational amplifier is coupled to the fourth transistor as avoltage follower, and wherein the first operational amplifier is furthercoupled to control a voltage at the first end of the resistor; and asecond operational amplifier operable to receive the error voltage,wherein the second operational amplifier is coupled to the fifthtransistor as a voltage follower, and wherein the second operationalamplifier is arranged to control a voltage at the second end of theresistor.
 8. The circuit of claim 7, wherein the resistor is anon-temperature-controlled resistor.
 9. The circuit of claim 7, whereinthe value of the adjustable reference current is based at least in parton the voltage at the first end of the resistor, the voltage at thesecond end of the resistor, and a value of the resistor.
 10. The circuitof claim 1, wherein the output circuit comprises: a first outputtransistor coupled to the reference current circuit and the comparisoncircuit such that the first output transistor conducts an outputreference current, wherein a value of the output reference current isequal to a sum of the constant reference current and the adjustablereference current; and a second output transistor operable to output thetemperature independent current limit, wherein the first outputtransistor and the second output transistor are coupled together as acurrent mirror.
 11. The circuit of claim 10, wherein the value of theoutput reference current is substantially equal to a value of theconstant reference current plus the value of the adjustable referencecurrent.
 12. The circuit of claim 1, wherein: a value of temperatureindependent current limit is equal to a first value when a voltage ofthe error signal is less than a voltage of the reference signal; thevalue of the temperature independent current limit changesproportionally with the voltage of the error signal when the voltage ofthe error signal is greater than the voltage of the reference signal andless than an upper threshold voltage; and the value of the temperatureindependent current limit is equal to a second value when the voltage ofthe error signal is greater than the upper threshold voltage, whereinthe second value is greater than the first value.
 13. The circuit ofclaim 1 incorporated within a controller for the power converter.
 14. Amethod for generating a temperature independent current limit toregulate an output of a power converter, the method comprising:generating a constant reference current; receiving an error signalrepresentative of a difference between an actual output value and adesired output value; receiving a reference signal representative of avalue of the error signal corresponding to a minimum current limit;generating an adjustable reference current, wherein a value of theadjustable reference current is based at least in part on a differencebetween the error signal and the reference signal; and generating theadjustable current limit based at least in part on the constantreference current and the adjustable reference current.
 15. The methodof claim 14, wherein the constant reference current is generated using atemperature trimmed current source.
 16. The method of claim 14, whereina minimum value of the temperature independent current limit is setbased at least in part on a value of the constant reference current, andwherein a maximum value of the temperature independent current limit isset based at least in part on the value of the constant referencecurrent and a maximum value of the adjustable reference current.
 17. Themethod of claim 16, wherein the maximum value of the temperatureindependent current limit is equal to at least four times the value ofthe minimum value of the temperature independent current limit.
 18. Themethod of claim 16, wherein the minimum value of the temperatureindependent current limit and the maximum value of the temperatureindependent current limit are temperature independent values.
 19. Themethod of claim 14, wherein generating the adjustable reference currentis performed using at least a resistor to convert the error signal intothe adjustable reference current.
 20. The method of claim 19, whereinthe resistor is a non-temperature-controlled resistor.
 21. A powerconverter comprising: an energy transfer element; a switch coupled tothe energy transfer element such that a current is conducted through theenergy transfer element and the power switch during an on time of theswitch; and a controller coupled to provide a drive signal to controlthe switch to regulate an output of the power converter, the drivesignal controlled based at least in part by a temperature independentcurrent limit, wherein the controller comprises: a feedback referencecircuit coupled to receive a feedback signal and coupled to generate anerror signal representative of a difference between an actual outputvalue and a desired output value; a reference current circuit operableto generate a constant reference current; a comparison circuit operableto: receive the error signal; receive a reference signal representativeof a value of the error signal corresponding to a minimum current limit;and generate an adjustable reference current, wherein a value of theadjustable reference current is variable based at least in part on adifference between the error signal and the reference signal; and anoutput circuit coupled to the reference current circuit and thecomparison circuit, the output circuit operable to generate thetemperature independent current limit based at least in part on theconstant reference current and the adjustable reference current.
 22. Thepower converter of claim 21, wherein the reference current circuitcomprises: a temperature trimmed current source operable to generate aconstant current; a first transistor coupled to the temperature trimmedcurrent source, wherein the first transistor is coupled to receive theconstant current from the temperature trimmed current source; and asecond transistor coupled to the first transistor as a current mirror,wherein the second transistor is coupled to receive the constantreference current.
 23. The power converter of claim 22, wherein thecomparison circuit comprises: a third transistor coupled to the firsttransistor as a current mirror; a fourth transistor coupled to the thirdtransistor; a fifth transistor coupled to the third transistor and thefourth transistor; a resistor coupled to the third transistor, thefourth transistor, and the fifth transistor; a first operationalamplifier operable to receive the reference signal, wherein the firstoperational amplifier is coupled to the fourth transistor as a voltagefollower, and wherein the first operational amplifier is further coupledto control a voltage at the first end of the resistor; and a secondoperational amplifier operable to receive the error voltage, wherein thesecond operational amplifier is coupled to the fifth transistor as avoltage follower, and wherein the second operational amplifier isfurther coupled to control a voltage at the second end of the resistor.24. The power converter of claim 23, wherein the resistor is anon-temperature-controlled resistor.
 25. The power converter of claim23, wherein the controller is operable to cause a change in the errorsignal that is proportional to a change in a value of the resistor. 26.The power converter of claim 23, wherein: in response to a decrease in avalue of the resistor, the controller is operable to cause an increasein the error signal; and in response to an increase in the value of theresistor, the controller is operable to cause a decrease in the errorsignal.